Linear voltage or current stabilizer LM317. Design, principle of operation of a pulse voltage stabilizer Adjustable pulse stabilizer

Rice. 2.4.11.

Operating principle of a PWM stabilizer. The switching frequency of the control transistor is constant. The ratio between the durations of the open and closed states of the control transistor changes. Two signals are supplied to the input of the comparing device (comparator), one of which U GPN comes from the sawtooth voltage generator, and the second from the output divider. Switching of the transistor will occur at the moment of equality of these signals. As Uin increases, KUout increases, which causes a decrease in the duration of the open state of the control transistor and a corresponding decrease in Un. Compared to relay stabilizers, PWM stabilizers are more complex and contain a larger number of elements.

Rice. 2.4.12.

In a stabilizer with PFM t and =const, and the frequency changes. The disadvantages of such a stabilizer are: the complexity of the control circuit, which ensures a change in frequency over a wide range; decrease in smoothing coefficient as frequency decreases. In stabilizers with PWM, you can select the optimal frequency at which the efficiency is greatest. In addition, in stabilizers with PFM and PWM, the output voltage ripple is less. In a relay stabilizer, U out~ fundamentally cannot be equal to zero, since periodic switching of the trigger in the control circuit is possible when U n changes in the range from U n.max to n.min.

Rice. 2.4.13.

In a pulse stabilizer with parallel connection of a transistor VT is open for t and =, U L U in, energy accumulates in the inductor, and the capacitor is discharged to the load. When the transistor is locked in the inductor, a self-inductive emf is induced. U out =U in +U L . Under the influence of this voltage, the diode opens and the capacitor is charged, U L =U out -U in. The DC component at the inductor is zero, so U in  = (U out - U in)(T - ) U out = U in  + U in - U in /(1 - ) = U in /( 1 - ) (2.4.7) This is a boost type stabilizer.

Rice. 2.4.14.

In an inverting stabilizer(Fig. 2.4.14) when VT is open for T, energy U L =U input is stored in the inductor, the capacitor is discharged to the load. When VT is closed, an EMF of reverse sign is induced in the throttle. U L =U out during duration T-T. The capacitor is charged from the inductor through an open diode. U in T=U out (T-T) U out =U in /(1-) (2.4.8). As the switching frequency of the control transistor increases, the relative duration of the processes of resorption of excess carriers in the base of the VT and the diode increases. This can lead to disruption of stable operation and transition to self-oscillation mode. Dynamic losses in the stabilizer elements increase and its efficiency decreases. Switching processes lead to a change in the shape of rectangular current and voltage pulses (the leading and trailing edges are delayed), but this is not so significant. What is significant is that VT experiences a large short-term overcurrent. When a control pulse is received at the base of a closed VT, opening it, Ik begins to increase, and the current through the blocking diode VD decreases. Since VD is still open, VT operates in short circuit mode and U in is applied to it and I to can be 5 to 10 times greater than I n. Thus, the inertia of real diodes is the main cause of switching overloads of control transistors. These overloads will be greater, the better the pulse properties of VT and the worse the performance of the diode. You have to choose a more powerful transistor, the current use of which will be low. To reduce overloads, current-limiting elements are introduced into the collector or emitter circuits. The introduction of an additional choke into the collector circuit is shown in Fig. 2.4.15.

Rice. 2.4.15.

L additional reduces the rate of increase of I k. R additional ensures that VD additional is locked at the moment the transistor VT opens. The choke discharge occurs when VT is closed through the diode VD add on R add. A two-winding inductor can be inserted into the collector or emitter circuit (Fig. 2.4.16).

Rice. 2.4.16.

The electromagnetic energy accumulated in L additional, when current flows through VT, returns back to the source when VT is closed. Compared to the previous case, the efficiency of the stabilizer increases due to the elimination of power losses in R add. When current flows through VD additional U ke.max =U in +U in W 1 /W 2. To reduce Uke.max, the ratio between W 1 and W 2 should be W 2 (5 10) W 1. In this case, the voltage amplitude on the closed diode U additional = (5 10) U input. In order to reduce U kn, t on and I ke0, the adjustable transistor is turned off by connecting to the base-emitter transition of the source U zap (Fig. 2.4.17a).

Rice. 2.4.17

When VT1 is open, VT2 is closed, C1 is charged by the base current I b1. When VT2 is unlocked, U c1 closes VT1. U c1 can vary depending on U input, U c1 is discharged to R 1. Therefore, instead of R 1, a zener diode or diodes are turned on in the forward direction (Fig. 2.4.17b). Although switching stabilizers are more economical than continuous ones, they have some disadvantages, the main of which are: 1) increased value of the output voltage ripple coefficient (for relays up to 10-20%, with PWM - 0.1-1%); 2) high dynamic internal resistance, that is, a falling external characteristic; 3) large interference created by the stabilizer, to reduce which additional filters are included at the input and output. This determines their scope of application: in power supply devices with a constant current load of significant power, where low weight and dimensions are required, but significant pulsations of U out are allowed. Currently, three types of integrated circuits (ICs) of pulse stabilizers are produced: 1) boost-type pulse stabilizers, powered by low input voltage from 2 to 12V, with minimal power dissipation and a built-in field-effect transistor (series of stabilizers 1446PN1, 1446PN2, 1446PN3); 2) universal low-power ICs that can be used to build a wide variety of switching stabilizer circuits (for example, 142EP1 or 1156EU1); 3) complete stabilizers, including a control circuit and a power transistor for a current of up to 10A (for example, 1155EU1). Table 1 shows the main characteristics of IC pulse stabilizers of these three groups. Step-up switching stabilizers 1446PN1, 1446 PN2 and 1446PN3 are designed to work with low input voltage and a fixed output voltage of +5 or +12V. The efficiency of such stabilizers reaches 88%, and the operating frequency is up to 170 kHz. At low output power, an internal field-effect transistor is used as a key element. To power powerful loads, it is necessary to use an additional bipolar or field-effect transistor. The main application of such ICs is in uninterruptible power supplies for individual computer boards, when powering measuring instruments from galvanic cells, and in portable communication devices.

Table 1 Main characteristics of ICs for controlling switching stabilizers

The most universal are the ICs of the second group, which are essentially a set of elements for constructing pulse stabilizers of various types. Of these microcircuits, the most advanced is the IC type 1156EU1, a simplified block diagram of which is shown in Fig. 2.4.18. The microcircuit is a set of standard switching stabilizer blocks located on one chip. The IC includes the following components and blocks: reference voltage source 1.25V; operational amplifier with a bias voltage of 4 mV, a gain of more than 200 thousand, a slew rate of 0.6 V/μs; pulse width modulator, including a master oscillator, a comparator, an “AND” circuit and an RS trigger; key transistor with driver (pre-amplifier); power diode with forward current 1A and reverse voltage 40V.

Rice. 2.4.18.

The microcircuit can control an external bipolar or field-effect transistor if an output current greater than 1.5A and a voltage greater than 40V are required. IC 142EP1 is used in a relay-type ISN circuit, the block diagram of which is shown in Fig. 2.4.19.

Rice. 2.4.19 ISN relay type.

FRP is a two-tier LC radio interference filter that weakens the voltage of radio interference introduced by the voltage stabilizer into the primary network during its operation. RE - a power transistor switch consisting of an IC type 286EP3 (a set of two powerful transistors), an additional boosting transistor VT and Dr, which limits the rate of increase of the current I to the transistor VT. SF - (VD, L and C), a filter that integrates a sequence of unipolar pulses. VF is a high-frequency filter that additionally attenuates the voltage of high-frequency ripples of the load current. Ultrasound is a protection device that provides protection against overloads (transistor protection). A reference voltage is supplied to one of the inputs of the differential UPT, and a voltage from the divider equal to the reference voltage is supplied to the other input. The mismatch signal is supplied to the Schmidt trigger through the emitter follower of the EP. At its output, unipolar pulses are generated, the duration of which varies depending on the UPT signal. These pulses control the PC parallel switch, which opens or closes the RE transistor.

The operation of almost any electronic circuit requires the presence of one or more constant voltage sources, and in the vast majority of cases a stabilized voltage is used. Stabilized power supplies use either linear or switching stabilizers. Each type of converter has its own advantages and, accordingly, its own niche in power supply circuits. The undoubted advantages of switching stabilizers include higher efficiency values, the ability to obtain high output current values ​​and high efficiency with a large difference between the input and output voltages.

The operating principle of a buck pulse stabilizer

Figure 1 shows a simplified diagram of the power section of the IPSN.

Rice. 1.

Field effect transistor VT performs high-frequency current switching. In pulse stabilizers, the transistor operates in switching mode, that is, it can be in one of two stable states: full conduction and cutoff. Accordingly, the operation of the IPSN consists of two alternating phases - the energy pumping phase (when the VT transistor is open) and the discharge phase (when the transistor is closed). The operation of the IPSN is illustrated in Figure 2.

Rice. 2. Operating principle of IPSN: a) pumping phase; b) discharge phase; c) timing diagrams

The energy pumping phase continues throughout the time interval T I. During this time, the switch is closed and conducts current I VT. Next, the current passes through the inductor L to the load R, shunted by the output capacitor C OUT. In the first part of the phase, the capacitor supplies current I C to the load, and in the second half, it takes part of the current I L from the load. The magnitude of the current I L continuously increases, and energy is accumulated in the inductor L, and in the second part of the phase - on the capacitor C OUT. The voltage across the diode V D is equal to U IN (minus the voltage drop across the open transistor), and the diode is closed during this phase - no current flows through it. The current I R flowing through the load R is constant (the difference I L - I C), accordingly, the voltage U OUT at the output is also constant.

The discharge phase occurs during the time T P: the switch is open and no current flows through it. It is known that the current flowing through the inductor cannot change instantly. The current IL, constantly decreasing, flows through the load and closes through the diode V D. In the first part of this phase, the capacitor C OUT continues to accumulate energy, taking part of the current I L from the load. In the second half of the discharge phase, the capacitor also begins to supply current to the load. During this phase, the current I R flowing through the load is also constant. Therefore, the output voltage is also stable.

Main settings

First of all, we note that according to their functional design, they distinguish between IPSN with adjustable and fixed output voltage. Typical circuits for switching on both types of IPSN are presented in Figure 3. The difference between them is that in the first case, the resistor divider, which determines the value of the output voltage, is located outside the integrated circuit, and in the second, inside. Accordingly, in the first case, the value of the output voltage is set by the user, and in the second, it is set during the manufacture of the microcircuit.

Rice. 3. Typical switching circuit for IPSN: a) with adjustable and b) with fixed output voltage

The most important parameters of IPSN include:

• Range of permissible input voltage values ​​U IN_MIN…U IN_MAX.
• The maximum value of the output current (load current) I OUT_MAX.
• Nominal value of the output voltage U OUT (for IPSN with a fixed output voltage value) or range of output voltage values ​​U OUT_MIN ...U OUT_MAX (for IPSN with an adjustable output voltage value). Often reference materials indicate that the maximum value of the output voltage U OUT_MAX is equal to the maximum value of the input voltage U IN_MAX. In reality this is not entirely true. In any case, the output voltage is less than the input voltage, at least by the amount of voltage drop across the key transistor U DROP. With an output current value equal to, for example, 3A, the value of U DROP will be 0.1...1.0V (depending on the selected IPSN microcircuit). Approximate equality of U OUT_MAX and U IN_MAX is possible only at very low load current values. Note also that the process of stabilizing the output voltage itself involves a loss of several percent of the input voltage. The declared equality of U OUT_MAX and U IN_MAX should be understood only in the sense that there are no other reasons for reducing U OUT_MAX other than those indicated above in a specific product (in particular, there are no obvious restrictions on the maximum value of the fill factor D). The value of the feedback voltage U FB is usually indicated as U OUT_MIN. In reality, U OUT_MIN should always be several percent higher (for the same stabilization reasons).
• Accuracy of output voltage setting. Set as a percentage. It makes sense only in the case of IPSN with a fixed output voltage value, since in this case the voltage divider resistors are located inside the microcircuit, and their accuracy is a parameter controlled during manufacturing. In the case of IPSN with an adjustable output voltage value, the parameter loses its meaning, since the accuracy of the divider resistors is selected by the user. In this case, we can only talk about the magnitude of the output voltage fluctuations relative to a certain average value (the accuracy of the feedback signal). Let us recall that in any case, this parameter for switching voltage stabilizers is 3...5 times worse compared to linear stabilizers.
• Voltage drop across open transistor R DS_ON. As already noted, this parameter is associated with an inevitable decrease in the output voltage relative to the input voltage. But something else is more important - the higher the resistance value of the open channel, the more energy is dissipated in the form of heat. For modern IPSN microcircuits, values ​​up to 300 mOhm are a good value. Higher values ​​are typical for chips developed at least five years ago. Note also that the value of R DS_ON is not a constant, but depends on the value of the output current I OUT.
• Duty cycle duration T and switching frequency F SW. The duration of the working cycle T is determined as the sum of the intervals T I (pulse duration) and T P (pause duration). Accordingly, the frequency F SW is the reciprocal of the operating cycle duration. For some part of the IPSN, the switching frequency is a constant value determined by the internal elements of the integrated circuit. For another part of the IPSN, the switching frequency is set by external elements (usually an external RC circuit), in this case the range of permissible frequencies F SW_MIN ... F SW_MAX is determined. A higher switching frequency allows the use of chokes with a lower inductance value, which has a positive effect on both the dimensions of the product and its price. Most ISPS use PWM control, that is, the T value is constant, and during the stabilization process the T I value is adjusted. Pulse frequency modulation (PFM control) is used much less frequently. In this case, the value of T I is constant, and stabilization is carried out by changing the duration of the pause T P. Thus, the values ​​of T and, accordingly, F SW become variable. In reference materials in this case, as a rule, a frequency is set corresponding to a duty cycle equal to 2. Note that the frequency range F SW_MIN ...F SW_MAX of an adjustable frequency should be distinguished from the tolerance gate for a fixed frequency, since the tolerance value is often indicated in reference materials manufacturer.
• Duty factor D, which is equal to the percentage
  the ratio of T I to T. Reference materials often indicate “up to 100%”. Obviously, this is an exaggeration, since if the key transistor is constantly open, then there is no stabilization process. In most models released on the market before approximately 2005, due to a number of technological limitations, the value of this coefficient was limited above 90%. In modern IPSN models, most of these limitations have been overcome, but the phrase “up to 100%” should not be taken literally.
• Efficiency factor (or efficiency). As is known, for linear stabilizers (fundamentally step-down) this is the percentage ratio of the output voltage to the input, since the values ​​of the input and output current are almost equal. For switching stabilizers, the input and output currents can differ significantly, so the percentage ratio of output power to input power is taken as efficiency. Strictly speaking, for the same IPSN microcircuit, the value of this coefficient can differ significantly depending on the ratio of the input and output voltages, the amount of current in the load and the switching frequency. For most IPSN, maximum efficiency is achieved at a load current value of the order of 20...30% of the maximum permissible value, so the numerical value is not very informative. It is more advisable to use the dependence graphs that are provided in the manufacturer’s reference materials. Figure 4 shows efficiency graphs for a stabilizer as an example. . Obviously, using a high-voltage stabilizer at low actual input voltage values ​​is not a good solution, since the efficiency value drops significantly as the load current approaches its maximum value. The second group of graphs illustrates the more preferable mode, since the efficiency value weakly depends on fluctuations in the output current. The criterion for the correct choice of a converter is not so much the numerical value of the efficiency, but rather the smoothness of the graph of the function of the current in the load (the absence of a “blockage” in the region of high currents).

Rice. 4.

The given list does not exhaust the entire list of IPSN parameters. Less significant parameters can be found in the literature.

Special Features
pulse voltage stabilizers

In most cases, IPSN have a number of additional functions that expand the possibilities of their practical application. The most common are the following:

• The “On/Off” or “Shutdown” load shutdown input allows you to open the key transistor and thus disconnect the voltage from the load. As a rule, it is used for remote control of a group of stabilizers, implementing a certain algorithm for applying and turning off individual voltages in the power supply system. In addition, it can be used as an input for emergency power off in case of an emergency.
• Normal state output “Power Good” is a generalizing output signal confirming that the IPSN is in normal operating condition. The active signal level is formed after the completion of transient processes from the supply of input voltage and, as a rule, is used either as a sign of the serviceability of the ISPN, or to trigger the following ISPN in serial power supply systems. The reasons why this signal can be reset: the input voltage drops below a certain level, the output voltage goes beyond a certain range, the load is turned off by the Shutdown signal, the maximum current value in the load is exceeded (in particular, the fact of a short circuit), temperature shutdown of the load and some other. The factors that are taken into account when generating this signal depend on the specific IPSN model.
• The external synchronization pin “Sync” provides the ability to synchronize the internal oscillator with an external clock signal. Used to organize joint synchronization of several stabilizers in complex power supply systems. Note that the frequency of the external clock signal does not have to coincide with the natural frequency of the FSW, however, it must be within the permissible limits specified in the manufacturer’s materials.
• The Soft Start function provides a relatively slow increase in output voltage when voltage is applied to the input of the IPSN or when the Shutdown signal is turned on at the falling edge. This function allows you to reduce current surges in the load when the microcircuit is turned on. The operating parameters of the soft start circuit are most often fixed and determined by the internal components of the stabilizer. Some IPSN models have a special Soft Start output. In this case, the startup parameters are determined by the ratings of external elements (resistor, capacitor, RC circuit) connected to this pin.
• Temperature protection is designed to prevent chip failure if the crystal overheats. An increase in the temperature of the crystal (regardless of the reason) above a certain level triggers a protective mechanism - a decrease in the current in the load or its complete shutdown. This prevents further rise in die temperature and damage to the chip. Returning the circuit to voltage stabilization mode is possible only after the microcircuit has cooled. Note that temperature protection is implemented in the vast majority of modern IPSN microcircuits, but a separate indication of this particular condition is not provided. The engineer will have to guess for himself that the reason for the load shutdown is precisely the operation of the temperature protection.
• Current protection consists of either limiting the amount of current flowing through the load or disconnecting the load. The protection is triggered if the load resistance is too low (for example, there is a short circuit) and the current exceeds a certain threshold value, which can lead to failure of the microcircuit. As in the previous case, diagnosing this condition is the concern of the engineer.

One last note regarding the parameters and functions of the IPSN. In Figures 1 and 2 there is a discharge diode V D. In fairly old stabilizers, this diode is implemented precisely as an external silicon diode. The disadvantage of this circuit solution was the high voltage drop (approximately 0.6 V) across the diode in the open state. Later designs used a Schottky diode, which had a voltage drop of approximately 0.3 V. In the last five years, designs have used these solutions only for high-voltage converters. In most modern products, the discharge diode is made in the form of an internal field-effect transistor operating in antiphase with the key transistor. In this case, the voltage drop is determined by the resistance of the open channel and at low load currents gives an additional gain. Stabilizers using this circuit design are called synchronous. Please note that the ability to operate from an external clock signal and the term “synchronous” are not related in any way.

with low input voltage

Considering the fact that in the STMicroelectronics range there are approximately 70 types of IPSN with a built-in key transistor, it makes sense to systematize all the diversity. If we take as a criterion a parameter such as the maximum value of the input voltage, then four groups can be distinguished:

1. IPSN with low input voltage (6 V or less);

2. IPSN with input voltage 10…28 V;

3. IPSN with input voltage 36…38 V;

4. IPSN with high input voltage (46 V and above).

The parameters of stabilizers of the first group are given in Table 1.

Table 1. IPSN with low input voltage

Back in 2005, the line of stabilizers of this type was incomplete. It was limited to microcircuits. These microcircuits had good characteristics: high accuracy and efficiency, no restrictions on the duty cycle value, the ability to adjust the frequency when operating from an external clock signal, and an acceptable RDSON value. All this makes these products in demand today. A significant drawback is the low maximum output current. There were no stabilizers for load currents of 1 A and higher in the line of low-voltage IPSN from STMicroelectronics. Subsequently, this gap was eliminated: first, stabilizers for 1.5 and 2 A ( and ) appeared, and in recent years - for 3 and 4 A ( , And ). In addition to increasing the output current, the switching frequency has increased and the open channel resistance has decreased, which has a positive effect on the consumer properties of the final products. We also note the emergence of IPSN microcircuits with a fixed output voltage ( and ) - there are not very many such products in the STMicroelectronics line. The latest addition, with an RDSON value of 35 mOhm, is one of the best in the industry, which, combined with extensive functionality, promises good prospects for this product.

The main application area for products of this type is battery-powered mobile devices. A wide input voltage range ensures stable operation of the equipment at different battery charge levels, and high efficiency minimizes the conversion of input energy into heat. The latter circumstance determines the advantages of switching stabilizers over linear ones in this area of ​​user applications.

In general, this group of STMicroelectronics is developing quite dynamically - approximately half of the entire line has appeared on the market in the last 3-4 years.

Switching buck stabilizers
with input voltage 10…28 V

The parameters of the converters of this group are given in Table 2.

Table 2. IPSN with input voltage 10…28 V

Eight years ago this group was represented only by microcircuits , and with input voltage up to 11 V. The range from 16 to 28 V remained empty. Of all the listed modifications, only , but the parameters of this IPSN poorly correspond to modern requirements. We can assume that during this time the nomenclature of the group under consideration has been completely updated.

Currently, the base of this group is microcircuits . This line is designed for the entire range of load currents from 0.7 to 4 A, provides a full set of special functions, the switching frequency is adjustable within a fairly wide range, there are no restrictions on the duty cycle, the efficiency and open-channel resistance values ​​meet modern requirements. There are two significant disadvantages in this series. Firstly, there is no built-in discharge diode (except for microcircuits with the D suffix). The accuracy of output voltage regulation is quite high (2%), but the presence of three or more external elements in the feedback compensation circuit cannot be considered an advantage. The microcircuits differ from the L598x series only in a different input voltage range, but the circuit design, and, consequently, the advantages and disadvantages are similar to the L598x family. As an example, Figure 5 shows a typical connection circuit for a three-amp microcircuit. There is also a discharge diode D and compensation circuit elements R4, C4 and C5. The F SW and SYNCH inputs remain free, therefore, the converter operates from an internal oscillator with the default frequency F SW.

Circuits of homemade pulse DC-DC voltage converters using transistors, seven examples.

Due to their high efficiency, switching voltage stabilizers have recently become increasingly widespread, although they are usually more complex and contain a larger number of elements.

Since in thermal energy Only a small fraction of the energy supplied to the switching stabilizer is converted, its output transistors heat up less, therefore, by reducing the area of ​​heat sinks, the weight and size of the device are reduced.

A noticeable disadvantage of switching stabilizers is the presence of high-frequency ripples at the output, which significantly narrows the scope of their practical use - most often switching stabilizers are used to power devices on digital microcircuits.

Step-down switching voltage stabilizer

A stabilizer with an output voltage lower than the input voltage can be assembled using three transistors (Fig. 1), two of which (VT1, VT2) form a key regulatory element, and the third (VT3) is an amplifier of the mismatch signal.

Rice. 1. Circuit of a pulse voltage stabilizer with an efficiency of 84%.

The device operates in self-oscillating mode. The positive feedback voltage from the collector of the composite transistor VT1 through capacitor C2 enters the base circuit of transistor VT2.

The comparison element and mismatch signal amplifier is a cascade based on the VTZ transistor. Its emitter is connected to the reference voltage source - zener diode VD2, and the base - to the output voltage divider R5 - R7.

In pulse stabilizers, the regulating element operates in switch mode, so the output voltage is regulated by changing the duty cycle of the switch.

Turning on/off transistor VT1 based on the signal from transistor VTZ is controlled by transistor VT2. At the moments when transistor VT1 is open, electromagnetic energy is stored in inductor L1, due to the flow of load current.

After the transistor closes, the stored energy is transferred to the load through the diode VD1. The ripples in the output voltage of the stabilizer are smoothed out by filter L1, SZ.

The characteristics of the stabilizer are entirely determined by the properties of the transistor VT1 and diode VD1, the speed of which should be maximum. With an input voltage of 24 V, output voltage of 15 V and a load current of 1 A, the measured efficiency value was 84%.

Choke L1 has 100 turns of wire with a diameter of 0.63 mm on a K26x16x12 ferrite ring with a magnetic permeability of 100. Its inductance at a bias current of 1 A is about 1 mH.

Step-down DC-DC voltage converter to +5V

The circuit of a simple switching stabilizer is shown in Fig. 2. Chokes L1 and L2 are wound on plastic frames placed in armored magnetic cores B22 made of M2000NM ferrite.

Choke L1 contains 18 turns of a harness of 7 wires PEV-1 0.35. A 0.8 mm thick gasket is inserted between the cups of its magnetic circuit.

The active resistance of the inductor winding L1 is 27 mOhm. Choke L2 has 9 turns of a harness of 10 wires PEV-1 0.35. The gap between its cups is 0.2 mm, the active resistance of the winding is 13 mOhm.

Gaskets can be made of rigid heat-resistant material - textolite, mica, electrical cardboard. The screw holding the magnetic circuit cups together must be made of non-magnetic material.

Rice. 2. Circuit of a simple key voltage stabilizer with an efficiency of 60%.

To set up the stabilizer, a load with a resistance of 5...7 Ohms and a power of 10 W is connected to its output. By selecting resistor R7, the rated output voltage is set, then the load current is increased to 3 A and, by selecting the size of capacitor C4, the generation frequency is set (approximately 18...20 kHz) at which high-frequency voltage surges on capacitor SZ are minimal.

The output voltage of the stabilizer can be increased to 8...10V by increasing the value of resistor R7 and setting a new operating frequency. In this case, the power dissipated by the VTZ transistor will also increase.

In switching stabilizer circuits, it is advisable to use electrolytic capacitors K52-1. The required capacitance value is obtained by connecting capacitors in parallel.

Main technical characteristics:

• Input voltage, V - 15...25.
• Output voltage, V - 5.
• Maximum load current, A - 4.
• Output voltage ripple at a load current of 4 A over the entire range of input voltages, mV, no more than 50.
• Efficiency, %, not lower than 60.
• Operating frequency at an input voltage of 20 b and a load current of 3A, kHz - 20.

An improved version of the +5V switching stabilizer

In comparison with the previous version of the pulse stabilizer, the new design of A. A. Mironov (Fig. 3) has improved and improved such characteristics as efficiency, stability of the output voltage, duration and nature of the transient process when exposed to a pulse load.

Rice. 3. Circuit of a pulse voltage stabilizer.

It turned out that when the prototype operates (Fig. 2), a so-called through current occurs through the composite switch transistor. This current appears at those moments when, based on a signal from the comparison node, the key transistor opens, but the switching diode has not yet had time to close. The presence of such a current causes additional heating losses of the transistor and diode and reduces the efficiency of the device.

Another drawback is the significant ripple of the output voltage at a load current close to the limit. To combat ripples, an additional output LC filter (L2, C5) was introduced into the stabilizer (Fig. 2).

The instability of the output voltage from changes in load current can only be reduced by reducing the active resistance of inductor L2.

Improving the dynamics of the transient process (in particular, reducing its duration) is associated with the need to reduce the inductance of the inductor, but this will inevitably increase the output voltage ripple.

Therefore, it turned out to be advisable to eliminate this output filter, and increase the capacitance of capacitor C2 by 5... 10 times (by parallel connecting several capacitors into a battery).

Circuit R2, C2 in the original stabilizer (Fig. 6.2) practically does not change the duration of the output current decline, so it can be removed (short circuit resistor R2), and the resistance of resistor R3 can be increased to 820 Ohms.

But then, when the input voltage increases from 15 6 to 25 6, the current flowing through resistor R3 (in the original device) will increase by 1.7 times, and the power dissipation will increase by 3 times (up to 0.7 W).

By connecting the lower output of resistor R3 (in the diagram of the modified stabilizer this is resistor R2) to the positive terminal of capacitor C2, this effect can be weakened, but at the same time the resistance of R2 (Fig. 3) should be reduced to 620 Ohms.

One of the effective ways to combat through current is to increase the rise time of the current through the opened key transistor.

Then, when the transistor is fully opened, the current through the diode VD1 will decrease to almost zero. This can be achieved if the shape of the current through the key transistor is close to triangular.

As calculations show, to obtain such a current shape, the inductance of storage choke L1 should not exceed 30 μH.

Another way is to use a faster switching diode VD1, for example, KD219B (with a Schottky barrier). Such diodes have higher operating speed and lower voltage drop at the same value of forward current compared to conventional silicon high-frequency diodes. Capacitor C2 type K52-1.

Improved device parameters can also be obtained by changing the operating mode of the key transistor. Features of work powerful transistor The advantage of the original and improved stabilizers is that it operates in an active mode, rather than in a saturated mode, and therefore has a high current transfer coefficient and closes quickly.

However, due to the increased voltage across it in the open state, the power dissipation is 1.5...2 times higher than the minimum achievable value.

You can reduce the voltage on the key transistor by applying a positive (relative to the positive power wire) bias voltage to the emitter of transistor VT2 (see Fig. 3).

The required value of the bias voltage is selected when setting up the stabilizer. If it is powered by a rectifier connected to a mains transformer, then a separate winding on the transformer can be provided to obtain the bias voltage. However, the bias voltage will change along with the network voltage.

Converter circuit with stable bias voltage

To obtain a stable bias voltage, the stabilizer must be modified (Fig. 4), and the inductor must be turned into transformer T1 by winding an additional winding II. When the key transistor is closed and the diode VD1 is open, the voltage on winding I is determined from the expression: U1=UBыx + U VD1.

Since the voltage at the output and at the diode changes slightly at this time, regardless of the value of the input voltage on winding II, the voltage is almost stable. After rectification, it is supplied to the emitter of transistor VT2 (and VT1).

Rice. 4. Scheme of a modified pulse voltage stabilizer.

Heating losses decreased in the first version of the modified stabilizer by 14.7%, and in the second - by 24.2%, which allows them to operate at a load current of up to 4 A without installing a key transistor on the heat sink.

In the stabilizer of option 1 (Fig. 3), the inductor L1 contains 11 turns, wound with a bundle of eight PEV-1 0.35 wires. The winding is placed in an armored magnetic core B22 made of 2000NM ferrite.

Between the cups you need to lay a 0.25 mm thick textolite gasket. In the stabilizer of option 2 (Fig. 4), transformer T1 is formed by winding two turns of PEV-1 0.35 wire over the inductor coil L1.

Instead of a germanium diode D310, you can use a silicon diode, for example, KD212A or KD212B, and the number of turns of winding II must be increased to three.

DC voltage stabilizer with PWM

A stabilizer with pulse-width control (Fig. 5) is close in principle to the stabilizer described in, but, unlike it, it has two feedback circuits connected in such a way that the key element closes when the load voltage exceeds or the current increases , consumed by the load.

When power is applied to the input of the device, the current flowing through resistor R3 opens the key element formed by transistors VT.1, VT2, as a result of which a current appears in the circuit transistor VT1 - inductor L1 - load - resistor R9. Capacitor C4 is charged and energy is accumulated in inductor L1.

If the load resistance is large enough, then the voltage across it reaches 12 B, and the zener diode VD4 opens. This leads to the opening of transistors VT5, VTZ and the closing of the key element, and thanks to the presence of the diode VD3, inductor L1 transfers the accumulated energy to the load.

Rice. 5. Stabilizer circuit with pulse-width control with efficiency up to 89%.

Stabilizer technical characteristics:

• Input voltage - 15...25 V.
• Output voltage - 12 V.
• Rated loading current is 1 A.
• Output voltage ripple at a load current of 1 A is 0.2 V. Efficiency (at UBX = 18 6, IN = 1 A) is 89%.
• Current consumption at UBX=18 V in load circuit closure mode is 0.4 A.
• Output short circuit current (at UBX =18 6) - 2.5 A.

As the current through the inductor decreases and capacitor C4 discharges, the voltage across the load will also decrease, which will lead to the closing of transistors VT5, VTZ and the opening of the key element. Next, the stabilizer operation process is repeated.

Capacitor C3, which reduces the frequency of the oscillatory process, increases the efficiency of the stabilizer.

With low load resistance, the oscillatory process in the stabilizer occurs differently. An increase in load current leads to an increase in the voltage drop across resistor R9, opening of transistor VT4 and closing of the key element.

In all operating modes of the stabilizer, the current it consumes is less than the load current. Transistor VT1 should be installed on a heat sink measuring 40x25 mm.

Choke L1 consists of 20 turns of a bundle of three PEV-2 0.47 wires, placed in a cup magnetic core B22 made of 1500NMZ ferrite. The magnetic core has a gap 0.5 mm thick made of non-magnetic material.

The stabilizer can be easily adjusted to a different output voltage and load current. The output voltage is set by choosing the type of zener diode VD4, and the maximum load current is set by a proportional change in the resistance of resistor R9 or by supplying a small current to the base of transistor VT4 from a separate parametric stabilizer through a variable resistor.

To reduce the level of output voltage ripple, it is advisable to use an LC filter similar to that used in the circuit in Fig. 2.

Switching voltage stabilizer with conversion efficiency 69...72%

The switching voltage stabilizer (Fig. 6) consists of a trigger unit (R3, VD1, VT1, VD2), a reference voltage source and a comparison device (DD1.1, R1), a direct current amplifier (VT2, DD1.2, VT5), a transistor switch (VTZ, VT4), an inductive energy storage device with a switching diode (VD3, L2) and filters - input (L1, C1, C2) and output (C4, C5, L3, C6). The switching frequency of the inductive energy storage device, depending on the load current, is in the range of 1.3...48 kHz.

Rice. 6. Circuit of a pulse voltage stabilizer with a conversion efficiency of 69...72%.

All inductors L1 - L3 are identical and are wound in B20 armored magnetic cores made of 2000NM ferrite with a gap between the cups of about 0.2 mm.

The rated output voltage is 5 V when the input voltage changes from 8 to 60 b and the conversion efficiency is 69...72%. Stabilization coefficient - 500.

The amplitude of the output voltage ripple at a load current of 0.7 A is no more than 5 mV. Output impedance - 20 mOhm. The maximum load current (without heat sinks for transistor VT4 and diode VD3) is 2 A.

Switching voltage stabilizer 12V

The switching voltage stabilizer (Fig. 6.7) with an input voltage of 20...25 V provides a stable output voltage of 12 V at a load current of 1.2 A.

Output ripple up to 2 mV. Due to its high efficiency, the device does not use heat sinks. The inductance of the inductor L1 is 470 μH.

Rice. 7. Circuit of a pulse voltage stabilizer with low ripple.

Transistor analogues: VS547 - KT3102A] VS548V - KT3102V. Approximate analogues of transistors BC807 - KT3107; BD244 - KT816.